Apparatus and method for controlling phase of signal

ABSTRACT

The present disclosure relates to an analog phase shifter for mitigating transmission losses. The analog phase shifter includes a multi-port network including an input port for inputting an RF signal and an output port for outputting a phase-changed RF signal. The analog phase shifter further includes a hybrid coupler configured to operably couple the input port and the output port to a plurality of load ports. The analog phase shifter additionally includes tunable reflective loads coupled to the hybrid coupler through the plurality of load ports. Load values of the tunable reflective loads are tuned by applying a plurality of independent voltages.

CROSS-REFERENCE TO RELATED APPLICATIONS AND CLAIM OF PRIORITY

This application is related to and claims priority to Russian PatentApplication No. 2016150642 filed on Dec. 22, 2016, and Korean PatentApplication No. 10-2017-0178071 filed on Dec. 22, 2017, the contents ofwhich are incorporated herein by reference.

TECHNICAL FIELD

The present disclosure generally relates to the field of microwaveanalog devices for phase shifting, and more particularly, to a microwaveanalog phase shifter based on tunable capacitances.

BACKGROUND

Main areas of the microwave analog phase shifters application areantenna arrays, such as focusing control for radar and communicationantennas and array beam steering, and use as microwave wireless powertransport (WPT) chargers. Furthermore, the microwave analog phaseshifters may be used as phase modulators for modulating a phase of aninput radio frequency (RF) signal.

General requirements for the microwave analog phase shifters are 360°phase shift, low insertion loss, low transmission coefficient ripple,compact size, printed circuit board (PCB) integrability as well as lowcost.

Tunable elements which are traditionally used in the microwave analogphase shifters may include semiconductors and dielectric varactors, andpiezoelectric capacitors. The most common design of the analog phaseshifters with tunable capacitances includes a 3-dB hybrid coupler (a2-branch line coupler, a rat-race coupler, a Lange coupler, or the like)with a tunable reflective load. The tunable reflective load usuallyincludes two tunable capacitors that are tuned simultaneously byapplying a varying direct current (DC) voltage. As a result, the lossesoccur that are caused by an inner resonance within a reflective loadstructure. The resonance losses lead to the high attenuation of an RFoutput signal and parasitic amplitude modulation. Such phenomena maydegrade system performance, that is, may reduce array gain (if used inantenna arrays), or may distort a signal spectrum (if used in a phasemodulator).

Known in the art is a phase shifter disclosed in U.S. Pat. No. 7,969,359B2. US' 359 B2 teaches a phase shifter including: a hybrid coupler whichis ground shielded and includes differential coplanar strip lines placedone on top of the other using different metal layers so that signalcoupling occurs vertically; and reflective terminations which areconnected to the hybrid coupler such that when the hybrid coupler isconnected to the reflective terminations, a phase shifter is formed, thereflective terminations each including a parallel LC circuit.Disadvantages of the known solution are high resonance losses, highripple, and high sensitivity of a phase response to voltage changes.

In addition, known in the art is a phase shifter disclosed in U.S. Pat.No. 6,710,679 B2 (U.S. '679). U.S. '679 teaches a 360° analog dielectricvaractor phase shifter, including a 180° analog rat-race ring phaseshifter, and a 180° digital switch line phase shifter, and the digitalphase shifter includes first and second microstrip lines connected toeach other through capacitors. One of the above microstrip lines servesas a 180° phase shift line and another microstrip line serves as areference lines. Disadvantages of the known solutions are very bulkydesign and high resonance losses.

Therefore, in designing a phase shifter, there is a demand for asolution having a simplified design and a reduced size with minimallosses to maintain the same amplitude ripple and a 360° phase range.

SUMMARY

To address the above-discussed deficiencies, it is a primary object toprovide at least the advantages described below. Accordingly, thepresent disclosure provides an apparatus and a method for reducingaverage losses and reducing a transmission ripple in an analog phaseshifter.

Another aspect of the present disclosure provides an apparatus and amethod for mitigating an inner resonance of reflective loads in ananalog phase shifter.

Furthermore, another aspect of the present disclosure provides anapparatus and a method for solving a problem of self-resonance ofreflective loads in an analog phase shifter.

According to an aspect of the present disclosure, there is provided ananalog phase shifter including: a multi-port network with an input portfor inputting an RF signal and an output port for outputting aphase-changed RF signal; a hybrid coupler configured to electricallycouple the input port and the output port to a plurality of load ports;and tunable reflective loads coupled to the hybrid coupler through theplurality of load ports, wherein load values of the tunable reflectiveloads are tuned by applying a plurality of independent voltages.

According to another aspect of the present disclosure, there is providedan operating method of an analog phase shifter including: receiving aninput on a phase shift; determining control voltage values correspondingto the input and applying the determined control voltage values tovarying loads.

The apparatus and the method according to various embodiments of thepresent disclosure can reduce average losses and ripple by mitigating aninner resonance of a microwave analog phase shifter.

The main technical effect achieved by the present disclosure is reducedand equalized transmission losses during the use of the proposed device.The above results in a higher data transmission rate provided forcommunication systems, higher operating range provided for radarsystems, and higher power transmission efficiency provided for WPTsystems. The proposed analog phase shifter has two independent controlchannels whereby each capacitance of the reflective load is tunedindependently. Hence, the inner resonance may be mitigated by asuccessive manipulation of both channels using a control program whichis optimized by the analysis of phase and losses functions of thecontrol voltages. Lowered losses lead to the higher efficiency of theproposed device, namely, the higher data transmission rate and longeroperating distances can be provided. Furthermore, a lowered ripplecauses no parasitic modulation. Hence, a higher signal integrity isprovided during the use of the disclosed apparatus and method.

The effects achieved by the present disclosure are not limited to theabove-mentioned effects, and other effects that are not mentioned hereincould be clearly understood by a person skilled in the art from thefollowing description.

Before undertaking the DETAILED DESCRIPTION below, it may beadvantageous to set forth definitions of certain words and phrases usedthroughout this patent document: the terms “include” and “comprise,” aswell as derivatives thereof, mean inclusion without limitation; the term“or,” is inclusive, meaning and/or; the phrases “associated with” and“associated therewith,” as well as derivatives thereof, may mean toinclude, be included within, interconnect with, contain, be containedwithin, connect to or with, couple to or with, be communicable with,cooperate with, interleave, juxtapose, be proximate to, be bound to orwith, have, have a property of, or the like; and the term “controller”means any device, system or part thereof that controls at least oneoperation, such a device may be implemented in hardware, firmware orsoftware, or some combination of at least two of the same. It should benoted that the functionality associated with any particular controllermay be centralized or distributed, whether locally or remotely.

Moreover, various functions described below can be implemented orsupported by one or more computer programs, each of which is formed fromcomputer readable program code and embodied in a computer readablemedium. The terms “application” and “program” refer to one or morecomputer programs, software components, sets of instructions,procedures, functions, objects, classes, instances, related data, or aportion thereof adapted for implementation in a suitable computerreadable program code. The phrase “computer readable program code”includes any type of computer code, including source code, object code,and executable code. The phrase “computer readable medium” includes anytype of medium capable of being accessed by a computer, such as readonly memory (ROM), random access memory (RAM), a hard disk drive, acompact disc (CD), a digital video disc (DVD), or any other type ofmemory. A “non-transitory” computer readable medium excludes wired,wireless, optical, or other communication links that transporttransitory electrical or other signals. A non-transitory computerreadable medium includes media where data can be permanently stored andmedia where data can be stored and later overwritten, such as arewritable optical disc or an erasable memory device.

Definitions for certain words and phrases are provided throughout thispatent document, those of ordinary skill in the art should understandthat in many, if not most instances, such definitions apply to prior, aswell as future uses of such defined words and phrases.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present disclosure and itsadvantages, reference is now made to the following description taken inconjunction with the accompanying drawings, in which like referencenumerals represent like parts:

FIG. 1 illustrates a view showing a structural diagram of a phaseshifter according to various embodiments of the present disclosure;

FIG. 2 illustrates a view showing a microstrip layout of a phase shifteraccording to various embodiments of the present disclosure;

FIG. 3 illustrates a view showing an electrical diagram of a tunablereflective load with open ended lines according to one preferredembodiment of the phase shifter proposed according to variousembodiments of the present disclosure;

FIG. 4 illustrates a view showing an electrical diagram of a tunablereflective load with shorted lines according to another preferredembodiment of the phase shifter proposed according to variousembodiments of the present disclosure;

FIG. 5 illustrates a view showing an electrical diagram of a tunablereflective load with shorted lines and an equalizing resistor accordingto still another preferred embodiment of the phase shifter proposedaccording to various embodiments of the present disclosure;

FIG. 6 illustrates a view showing a reflection coefficient magnitude mapfor an exemplary reflective load as a function of C₁, C₂ according tovarious embodiments of the present disclosure;

FIG. 7 illustrates a view showing a reflection coefficient magnitude foran exemplary reflective load as a function of two control voltages whenboth capacitances are tuned simultaneously according to variousembodiments of the present disclosure;

FIG. 8 illustrates a view showing a reflection coefficient magnitude foran exemplary reflective load as a function of two control voltages whenboth capacitances are tuned independently using an optimal tuning pathwith and without an equalizing resistor according to various embodimentsof the present disclosure;

FIG. 9 illustrates a view showing a measured transmission coefficientmagnitude for a full phase shifter prototype versus frequency andcontrol voltages according to various embodiments of the presentdisclosure; and

FIG. 10 illustrates a view showing a flowchart for controlling a phasein the phase shifter according to various embodiments of the presentdisclosure.

DETAILED DESCRIPTION

FIGS. 1 through 10, discussed below, and the various embodiments used todescribe the principles of the present disclosure in this patentdocument are by way of illustration only and should not be construed inany way to limit the scope of the disclosure. Those skilled in the artwill understand that the principles of the present disclosure may beimplemented in any suitably arranged system or device.

The terms used in the present disclosure are just for the purpose ofdescribing particular exemplary embodiments, and are not intended tolimit the scope of other embodiments. The singular forms are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. All of the terms used herein including technical orscientific terms may have the same meanings as those generallyunderstood by an ordinary skilled person in the related art. From amongthe terms used in the present disclosure, the terms defined in agenerally used dictionary should be interpreted as having the same orsimilar meanings as or to the contextual meanings of the relevanttechnology, and should not be interpreted as having ideal or exaggeratedmeanings unless they are clearly defined in the present disclosure.According to circumstances, even the terms defined in the presentdisclosure should not be interpreted as excluding the embodiments of thepresent disclosure.

In various embodiments of the present disclosure which will be describedhereinbelow, hardware-level approach methods will be described by way ofan example. However, since various embodiments of the present disclosureinclude technology using both hardware and software, various embodimentsof the present disclosure do not exclude software-based approachmethods.

The present disclosure relates to an apparatus and a method forcontrolling a reflective load in a phase shifter. Specifically, thepresent disclosure describes technology for independently tuning eachcapacitance of a reflective load in a phase shifter.

The terms used in the following description to indicate elements of theapparatus (a varactor, a microstrip, a coupler, a port, or the like) aremerely examples for convenience of explanation. Therefore, the presentdisclosure is not limited to the terms described below, and other termshaving the same technical meanings may be used.

In addition, the present disclosure will be described with variousembodiments using a hybrid coupler, but this is merely an example forconvenience of explanation. Various embodiments of the presentdisclosure may be easily modified and applied to other couplers (forexample, a rat-race coupler, a Lange coupler).

FIG. 1 shows a structural diagram of a phase shifter according tovarious embodiments of the present disclosure.

Referring to FIG. 1, the phase shifter 100 includes an RF input port102, an RF output port 104, a three-decibel (dB) hybrid coupler 106,tunable reflective loads 108 and 110, independent control channels 112and 114, DC ports 116 and 118, a processor 120, and a memory 130.

The RF input port 102 and the RF output port 104 may form a networkhaving two ports of the phase shifter, and the 3-dB hybrid coupler 106may electrically couple the RF input port 102 and the RF output port 104to load ports including two identical tunable reflective loads 108 and110. The tunable reflective loads 108 and 110 may have the samestructure, and the control channels 112 and 114 may be connected to thetunable reflective loads 108 and 110 to independently controlcapacitance of each of the reflective loads. The processor 120 may inputa voltage for phase control through the DC ports 116 and 118. The memory130 may store a table indicating a relationship between capacitance ofeach of the varactors which are tuned by voltage vales applied to the DCports 116 and 118, and a corresponding reflection coefficient.

Each reflective load ideally should exhibit as a perfect reflector. Thatis, each reflective load should have a constant magnitude of areflection coefficient close to 1 (0 dB) and a phase varying from 0 to360 degrees or from 0 to −360 degrees. To achieve such operationparameters, each reflective load contains at least two tunablecapacitors. The reflection coefficient phase may be changed by varyingthe capacitance of each capacitor. Unfortunately, each capacitor(semiconductor, dielectric varactor, etc.) at microwave frequencies hassome parasitic losses. Thus, the magnitude of the reflection coefficientalso varies during phase shifting. This parasitic amplitude modulation(ripple) is well known for simple reflective load structures and causesphase shifter transmission resonance losses when the two capacitors withthe reflective load achieve 360 degrees. Phase shifting using onecontrol voltage is performed, that is, two capacitors are tunedsimultaneously. The present disclosure aims at reducing averagereflective load losses and a parasitic ripple by a reflective loadstructure modification and implementation of an optimal algorithm for acapacitor control.

According to various embodiments of the present disclosure, the hybridcoupler may be implemented as a microstrip or coplanar two-branch linecoupler having two similar reflective loads connected to load ports ofthe coupler, and the load ports are coupled to the input port of thecoupler.

Each reflective load may consist of two similar varactor diodes(referred to as varactors) having coplanar transmission lines ormicrostrip of a predetermined length and characteristic impedancetherebetween as described in detail below. Each varactor defines avaractor section including the varactor and a segment of a tuningmicrostrip or a coplanar line terminated by a shorted circuit or an opencircuit on the end of the varactor. Each varactor includes its own DCcontrol port connected thereto through a DC filter. When the tuninglines of the varactor are open ended, a DC port may be connected tothese lines, while the transmission line between two varactors may beshorted through another DC filter. In the other case, when tuning linesare shorted, each varactor may be biased by a separate DC port connectedto the transmission line between two varactors. In this case, two DCports should be isolated by an additional blocking capacitance insertedinside a gap in the line. In other words, a blocking capacitor connectedin series with the transmission line should be provided. Two varactorsare tuned independently to achieve the best performance in terms of thetransmission losses.

When phase shifting using one control voltage is performed, i.e., bothcapacitors are tuned simultaneously, the present disclosure aims atreducing average reflective load losses and a parasitic ripple by areflective load structure modification and implementation of an optimalalgorithm for capacitor control.

Although FIG. 1 depicts the phase shifter including a network having twoports, a hybrid coupler having two load ports, two reflective loads, andtwo DC ports, the present disclosure is not limited thereto. The phaseshifter according to the present disclosure may include three or moreload ports, three or more input and output ports, three or morereflective loads, and three or more DC ports.

The analog phase shifter may further include a configuration forgenerating an RF signal although it is not illustrated. Specifically,the phase shifter may further include an amplifier, an oscillator, amultiplier, a mixer, a filter, a duplexer, an isolator, or etc.

FIG. 2 shows a microstrip layout of a phase shifter according to variousembodiments of the present disclosure.

Referring to FIG. 2, the microstrip phase shifter includes an inputmicrostrip line 201, an output microstrip line 202 with characteristicimpedance Z₀, and a two-branch line 3-dB hybrid coupler having two loadports 204, 205 configured to couple identical reflective loads 108, 110.The characteristic impedance is Z₀ for all ports of the 3-dB hybridcoupler. Next, each of the load ports 204, 205 of the coupler 203 isconnected to a reflective load input having the characteristic impedanceZ₀. Thus, the load ports 204, 205 and the reflective load inputs areformed so as to have the same microstrip part. Each reflective load 108,110 may further include one varactor 243, 244 with capacitance C_(var2)connected in parallel to the input microstrip line. The varactor 243,244 is connected to another varactor 241, 242 with capacitance C_(var1)through a quarter (¼) wavelength microstrip line, which is also referredto as a transmission line 245, 246. Each of the varactors 241, 242 and243, 244 may include corresponding tuning microstrip open circuit lines214, 215 and 212, 214 connected thereto in series to form a tunablevaractor section. These lines are used for achieving a desired phaseshift by each varactor section. A phase shift desired from each varactorsection is usually 180 degrees and tuning line lengths andcharacteristic impedance are chosen based on varactor tunability. Thecharacteristic impedance of the transmission lines 245, 246 is usuallychosen to match the reflective load with the coupler ports, i.e., toachieve the lowest losses and ripple. Herein, the above-describedtransmission lines may include an open transmission line or a shortedtransmission line, and may be desired to achieve a phase shift of 180degrees. Accordingly, the transmission lines may essentially have aninductive reactance connected to the varactors in series. The differencebetween the shorted transmission line and the open transmission line isa different length between the above-described lines to achieve a valueof the inductance desired for achieving the phase shift of 180 degrees.DC-pass filters 216, 217, 218, 219 are exploited to provide independentvaractors for controlling channels 112 and 114 through a voltage appliedto the two DC ports 208, 209. In this preferred embodiment, it should berealized that the filters are the same as a microwave stopband filtertuned to f₀, where f₀ is a center microwave frequency of an operatingbandwidth. It should be understood that other possible filters havingthe same operating characteristics can be used. To maintain a DC groundon the other varactors' contacts, a shorted quarter (¼) wavelength stub220 is used to ground the coupler 106 together with the reflective loads108, 110.

Although FIG. 2 has been described based on the phase shifter includinga network with two ports, a hybrid coupler with two load ports, tworeflective loads, and two DC ports, the present disclosure is notlimited thereto. The phase shifter according to the present disclosuremay include three or more load ports, three or more input and outputports, three or more reflective loads, and three or more DC ports.

The described design may be manufactured on standard microwavesubstrates, e.g., Teflon based substrates or alumina substrates, usingexisting technologies: PCB etching, vacuum deposition, low-temperatureco-fried ceramics (LTCC), etc. The microstrip layout may be directlyintegrated inside a microwave system layout as a part of it, or may beimplemented as a separate device with its own housing and connectors(for example, subminiature version A (SMA) connectors).

The presented exemplary layout includes two separated DC ports. As willbe described in detail below, the reflective load proposed through FIGS.3 to 5 may have three different structures. The diagram of eachreflective load according to FIGS. 3 to 5 may correspond to the layoutof the microstrip of the reflective load illustrated in FIG. 2.

FIG. 3 shows an electrical diagram of a tunable reflective load withopen ended lines according to one preferred embodiment of the phaseshifter proposed according to various embodiments of the presentdisclosure.

Referring to FIG. 3, the tunable reflective load 304 includes twosimilar tunable capacitances C_(var1,2) (for example, varactors 341,343) separated by a quarter (¼) wavelength transmission line 345 withcharacteristic impedance 2Z₀, where Z₀ is the aforementioned referencecharacteristic impedance.

Each varactor 341, 343 has similar tuning open circuit lines 314, 312with a length and characteristic impedance chosen to achieve a phaseshift of 180 degrees from each varactor section. This choice depends ona varactor tuning range (C_(max) and C_(min)) and a value of Z₀. Thedesign equation may be derived from the condition of 180 degrees of thephase shift:

$\begin{matrix}{{\left( {{- \frac{1}{2\;\pi\; f_{0}C_{\min}}} + X_{s}} \right)/\left( {2Z_{0}} \right)} = {{- 2}{Z_{0}/\left( {{- \frac{1}{2\;\pi\; f_{0}C_{\max}}} + X_{s}} \right)}}} & {{Equation}\mspace{14mu}(1)}\end{matrix}$

where X_(s) is a full reactance of the tuning open circuit line havingthe varactor's parasitic inductance, Z₀ is characteristic impedance, f₀is a central frequency, C_(max) is a maximal value of the tunablecapacitance, and C_(min) is a minimal value of the tunable capacitance.The length of the tuning open circuit line is usually between λ₀/4 andλ₀/2, where λ₀ is a transmission line wavelength at the centralfrequency f₀.

By connecting two DC ports 308, 309 to the tuning open circuit lines314, 312 through the corresponding DC-pass filters 316, 318, decouplingbetween the DC ports 308, 309 is achieved. In this case, thetransmission line 345 should be grounded through another DC-pass filter320 to maintain a reference DC voltage level on the varactors' contactsconnected to the transmission line 345.

FIG. 4 shows an electrical diagram of a tunable reflective load withshorted lines according to another preferred embodiment of the phaseshifter proposed according to various embodiments of the presentdisclosure.

Referring to FIG. 4, the tunable reflective load 404 includes the samemicrowave parts as in the embodiment according to FIG. 3. However,tuning lines 412, 414 are shorted. A length and characteristic impedanceof the tuning shorted lines 412, 414 are calculated from the Equation(1), where X_(s) is a full reactance of the tuning shorted line havingthe varactor's parasitic inductance. In this case, a DC voltage shouldbe applied to varactors 441, 443 connected to a transmission line 445through corresponding DC-pass filters 418, 416. Hence, to achieve the DCports 408, 409 decoupling, an additional DC blocking capacitor 450 isconnected in series with the transmission line 445. A reactance at themicrowave frequency f₀ of the blocking capacitor 450 is at least anorder of a magnitude less than Z₀.

FIG. 5 shows an electrical diagram of a tunable reflective load withshorted lines and an equalizing resistor according to still anotherpreferred embodiment of the phase shifter proposed according to variousembodiments of the present disclosure.

Referring to FIG. 5, the electrical diagram of the reflective load 504includes the same parts as in the embodiment according to FIG. 4.Additionally, the reflective load 504 includes a resistor 555 havingresistance R_(p), and the resistor 555 is connected in parallel with avaractor 541. Further, a matching open circuit stub or line 560 isconnected in parallel with a varactor 543. A value of R_(p) is chosen toequalize reflection losses according to equation presented below:

$\begin{matrix}{R_{p} = \frac{\left( {2Z_{0}} \right)^{2}}{R_{v}}} & {{Equation}\mspace{14mu}(2)}\end{matrix}$

where R_(v) is an active resistance of the varactor 541 at the microwavefrequency, and Z₀ is characteristic impedance.

The equalizing resistor 555 is connected in series with another DCblocking capacitor 551. A reactance on the microwave frequency f₀ of theblocking capacitor 551 is at least an order of a magnitude less thanR_(p). The open circuit line 560 is further used for transmission lossesequalization by adjusting the full reflective load reactance. A lengthand characteristic impedance of the open circuit line 560 is chosen froma resonance equation:

$\begin{matrix}{{{- \frac{1}{2\;\pi\; f_{0}C_{\max}}} + X_{s}} = {- X_{p}}} & {{Equation}\mspace{14mu}(3)}\end{matrix}$

where X_(p) is a reactance of the open circuit line 560, X_(s) is a fullreactance of the tuning shorted line having the varactor's parasiticinductance, and C_(max) is a maximal value of the tunable capacitance.

In one of the embodiments, the device structure is the same as in theprevious embodiments, and there is proposed a phase shifter additionallyincluding an equalizing resistor connected in parallel to a point whereone of the varactors is connected to a transmission line between thevaractors. Additionally, the structure may include a matching opencircuit stub or line which is connected in parallel to a point whereanother varactor (being closest to the coupler) is connected to thetransmission line between the varactors. Preferably, the equalizingresistor may be connected in series with another blocking capacitor. Thetwo varactors are tuned independently to achieve the best performance interms of the transmission losses.

Thus, in any of the above-described embodiments, each capacitance may becontrolled independently which results in that the reflective load maybe controlled more versatile, and the tuned capacitances may work inless heavy regime.

Referring to FIGS. 6 to 8, the advantage of the present disclosure isthat the inner resonance of the reflective load 108, 110 can beeffectively mitigated by the successive manipulation of the both controlchannels 112, 114. A reflection coefficient of the reflective load 108,110 is a two-dimensional function of C_(var1) and C_(var2).

FIG. 6 shows a reflection coefficient magnitude map for an exemplaryreflective load as a function of C₁, C₂ according to various embodimentsof the present disclosure.

Referring to FIG. 6, the map was constructed based on a simulation ofthe reflective load 404 with the following parameters Z₀=50 Ohm,C_(min)=0.15 pF, C_(max)=0.6 pF, R_(v)=4 Ohm at f₀=5.8 GHz for thecircuit according to the embodiment illustrated in FIG. 4. A resonancearea 600 is shown by the darkest area on the map.

During the capacitance tuning from the maximal value to the minimalvalue, the reflective load 404 changes the state thereof from the upperright corner of the map (point 601) to the lower left corner (point602). When all parameters are chosen according to the above Equations(1) to (3), the total phase shift during the movement along any pathbetween these two points is 360 degrees. It should be noted that eachpath has its own average losses and ripple. The idea is to choose thepath with minimal average losses and ripple, i.e. to avoid the resonancearea 600, where the average losses and ripple are maximal.

When a known algorithm is used, i.e., both C_(var1) and C_(var2) aretuned simultaneously, the circuit is tuned along the diagonal path(dashed line 603, a tuning direction is shown by an arrow). It isclearly shown in FIG. 6 that the circuit goes through or nearby theresonance area 600 during the tuning. On the other hand, when eachcapacitance is tuned independently, the resonance may be got round by,for example, the lower right corner (point 604) as depicted in FIG. 6 asthe optimal path (dotted lines 605 and 606, the tuning direction isshown by the arrow).

For some other cases, the tuning path may be formed as a quarter (¼) ofthe circle passing through the upper right point 601 and lower leftpoint 602 and above the lower right point 604 of the loss map (seedash-and-dotted curved line 607 on FIG. 6).

Specifically, as a non-limiting example of the control algorithm, analgorithm as described below may be used: C_(var1) is tuned from C_(max)to C_(min) (corresponding to the line 605 extending from point 601 topoint 604), while C_(var2) is maintained at C_(max), and then C_(var2)is tuned from C_(max) to C_(min) (corresponding to the line 606extending from point 604 to point 602), while C_(var1) is maintained atC_(min). Another way is to perform a phase shift according to, forexample, line 607 from 601 to 602 by tuning each of C_(var1) andC_(var2) independently.

It should be understood that any suitable means known in the art may beused to implement the above algorithm. As a non-limiting example, suchmeans includes a program code being stored in a computer readablemedium, and said program code includes instructions to realize avaractor DC control.

The varactor DC control may be implemented using, for example, astandard digital to analog converter (DAC). In this case, a digitalcontroller or another microprocessor device being used to control thephase shifter may be configured to send a digital code to the DAC input.The DAC may be further configured to form an output analog voltage to beapplied to the DC ports of the proposed phase shifter. For the most ofapplications, 8 bits per each DC control channel is enough for a precisephase control.

FIG. 7 shows a reflection coefficient magnitude for an exemplaryreflective load as a function of two control voltages when bothcapacitances are tuned simultaneously according to various embodimentsof the present disclosure.

Referring to FIG. 7, simulation results for the reflection coefficientmagnitude as a function of two DC control voltages during thesimultaneous tuning from C_(max) to C_(min) according to the knownalgorithm (path 603 shown in FIG. 6) are illustrated by curved line 703.In particular, at the initial time 701, control voltages U_(DC1),U_(DC2) being applied to the control DC ports are minimal, whileC_(var1), C_(var2) are maximal, and, at the finish time 702, controlvoltages U_(DC1), U_(DC2) are maximal, while C_(var1), C_(var2) areminimal.

FIG. 8 shows a reflection coefficient magnitude for an exemplaryreflective load as a function of two control voltages when bothcapacitances are tuned independently using an optimal tuning path withand without an equalizing resistor according to various embodiments ofthe present disclosure.

Referring to FIG. 8, the results simulated for the above proposedcontrol algorithm are presented by curves 803 and 830. In particular, atthe initial time 801, control voltages U_(DC1), U_(DC2) are minimal,while C_(var1), C_(var2) are maximal; at the intermediate time 804 (whentuning path illustrated in FIG. 6 changes the direction from line 605 toline 606 at point 604), U_(DC1) is maximal and U_(DC2) is minimal, whileC_(var1) is minimal and C_(var2) is maximal; and, at the finish time802, U_(DC1), U_(DC2) are maximal, while C_(var1), C_(var2) are minimal.

Average losses for the case without the equalizing resistor (see curve803) were improved by 0.7 dB and the ripple level was reduced from 1.6dB to 0.6 dB. For the case with the equalizing resistor according to theembodiment of FIG. 5 (see curve 830) the average losses were improved by0.35 dB and the ripple level was reduced from 1.6 dB to 0.1 dB.

Thus, in the present disclosure, the resonance losses are mitigated bychoosing an optimal control algorithm which results in that the averagereflection losses and a transmission ripple of the reflective load arereduced. To summarize, with respect to a ripple (for example, 0.5 dB) ofa pre-given acceptable value, an optimal control algorithm and/orcontrol program may provide the least average insertion losses of thephase shift. Herein, optimizing may be performed at the stage ofdevelopment, and this means that the optimal control algorithm and/orcontrol program for varactor tuning may be used when a phase shifterusing specific configurations (varactors, PCB, or etc.) is designed.Specifically, a path according to independent control may be one or morepaths, and each path may have its own average losses and ripple. Herein,determined control voltage values may be final voltage values forachieving a desired phase shift, and the path may indicate a pluralityof control voltage values continuously applied to the DC ports toachieve the final voltage values. Accordingly, the optimal controlalgorithm and/or control program may determine a plurality of paths forachieving certain control voltage values, and may determine these pathsas a candidate group for the control voltage values. However, each pathmay have its own average losses and ripple.

Thereafter, the optimal control algorithm and/or control program may bedetermined, and may be loaded into a controller memory of the phaseshifter at the stage of manufacturing. That is, the control algorithmand/or control program may be implemented through a controller or aprocessor. In addition, the control algorithm and/or control program maybe essentially based on a table showing a relationship between a phaseshift (from 0 to 360 degrees) and two DC voltages applied to two controlchannels of the phase shifter.

According to one embodiment, the controller or processor receives arequest for setting an output of the phase shifter to a value given foran output phase. The controller or processor may determine two controlvoltages corresponding to the closest phase corresponding to the desiredoutput phase using the above-mentioned table. Thereafter, the controlleror processor may set the two control voltages to the determined valuesby transmitting an appropriate digital code to a DAC having an outputconnected to the control ports of the phase shifter.

In another embodiment, the controller or processor may receive a valueon a phase shift which is used for beam steering according to a beamindex determined on a higher layer. The controller or the processor maydetermine two control voltage values for achieving the phase changeamount using the table showing the relationship between the phase shiftand two DC voltages applied to the control channels, which is stored inthe memory. Thereafter, the controller or processor may transmit signalsto the DAC having the output connected to the control ports of the phaseshifter, and may control the two control voltages based on thedetermined values.

The proposed phase shifter according to the embodiment shown in FIG. 2was tested under the following conditions: Rogers R04003C substrate,MA-COM MA46H120 varactor diodes, 1-12 DC voltage range, and 5.8 GHzcentral frequency. Measurements were done using the Agilent PNA-X vectornetwork analyzer.

FIG. 9 shows a measured transmission coefficient magnitude for a fullphase shifter prototype versus frequency and control voltages accordingto various embodiments of the present disclosure.

Referring to FIG. 9, the transmission losses are less than 2 dB for the360⁰ phase range, the ripple amounts are less than 0.7 dB, geometricalarea of the phase shifter is about 0.5×0.5 wavelength², and figure ofmerit equals to 211 deg./dB, which are sufficiently better parametersthan the corresponding parameters of the known systems. Comparing to thesimulated transmission losses for the solution of U.S. '679 estimatingabout 3-5 dB, the geometrical area is about 1×0.5 wavelength², and thefigure of merit estimates about 100-130 deg./dB with a sufficientlycomplicated design.

FIG. 10 shows a flowchart for controlling a phase in the phase shifteraccording to various embodiments of the present disclosure. Therespective operations in FIG. 10 may be performed by the controller orprocessor of the phase shifter.

Referring to FIG. 10, in operation 1001, the analog phase shifterreceives an input on a phase shift. For example, when the phase shifteris implemented as a part of a base station, the base station may receivefeedback on a preferred beam index from a terminal, and may determine avalue of a phase shift to be used for base station transmission beamsteering based on the received feedback. Accordingly, the controller orprocessor may receive the value of the phase shift as an input. Inanother example, when the phase shifter is implemented as a part of asatellite tracking receiver, the satellite tracking receiver may includean antenna interferometer using four antennas, and may track a motion ofa satellite under the auto control of the antenna interferometer. Inresponse to the motion of the satellite tracked under the auto control,a phase shift of the receiver may be used. Accordingly, the controlleror processor may receive a value of the phase shift according to thetracked motion of the satellite as an input.

In operation 1003, the analog phase shifter may determine correspondingcontrol voltage values. Specifically, the controller or the processormay determine control voltage values for achieving a phase change amountusing information on a relationship between a phase change amount of anRF signal and two DC voltages applied to control channels, which isstored in the memory. For example, the controller or processor receivesa phase change amount that is used for beam steering, and may determinecontrol voltage values for achieving the corresponding phase changeamount. In another example, the controller or the processor may receivea phase change amount that is used for tracking a satellite, and maydetermine corresponding control voltage values using the table. Thecontroller or processor may determine voltage values corresponding tothe phase change amount from among a plurality of candidates, andcombinations of voltage values included in the plurality of candidatesmay be defined in a region other than a region including combinations ofvoltage values causing the inner resonance of reflective loads.

In operation 1005, the analog phase shifter applies the determinedcontrol voltage values to varying loads. Specifically, the controller orprocessor may apply independent control voltage values corresponding tothe phase change amount to reflective loads through control channels.The independent control voltage values may be applied to a DAC connectedto control ports, and may be a signal having an appropriate digitalcode.

Although FIG. 10 depicts the phase shifter using two control voltages,the present disclosure is not limited thereto. The phase shifteraccording to the present disclosure may include three or more reflectiveloads and three or more independent control voltage values appliedthereto independently.

As a non-limiting example, the use of the proposed phase shifter in amicrowave radiating system is given below. One of the example is a WPTsystem where the proposed phase shifter is integrated inside an 8×8transmitting antenna array with a capability of autofocusing on areceiver. In that system, a method for generating a phase distributionover an array aperture may be implemented using the proposed phaseshifter.

Another exemplary system is a satellite tracking receiver where theproposed phase shifter is used for the phase adjustment during thesatellite tracking by an automatic control of a four-antennainterferometer.

Still another exemplary system is a controllable oscillator where theproposed phase shifter is used for the phase adjustment.

As described above, the proposed disclosure has numerous positiveeffects for RF and microwave systems such as a higher link budget andhigher signal quality. Hence, the proposed solution can beadvantageously used in the radar antenna arrays, base station antennas,satellite communication systems, controllable oscillators, radiofrequency integrated circuits (RFICs), etc. Furthermore, it is possibleto design the microwave analog phase shifters based on the tunablecapacitances (with operating frequencies up to 100 GHz) for the use ininternet of things (IoT) sensors, wireless fidelity (Wi-Fi) and mobilecommunication, long distance WPT systems, etc.

Methods according to embodiments stated in claims and/or specificationsof the present disclosure may be implemented in hardware, software, or acombination of hardware and software.

The software may be stored in a computer-readable storage medium. Thecomputer-readable storage medium stores at least one program (softwaremodule) including instructions that causes, when executed by at leastone processor in the electronic device, the electronic device to performthe method of the present disclosure.

The programs (software modules or software) may be stored innon-volatile memories including a random access memory and a flashmemory, a read only memory (ROM), an electrically erasable programmableread only memory (EEPROM), a magnetic disc storage device, a compactdisc-ROM (CD-ROM), digital versatile discs (DVDs) or other type opticalstorage devices, or a magnetic cassette. Alternatively, any combinationof some or all thereof may form a memory in which the program is stored.Further, a plurality of such memories may be included in the electronicdevice.

In addition, the programs may be stored in an attachable storage devicewhich may access the electronic device through communication networkssuch as the internet, intranet, local area network (LAN), wide LAN(WLAN), and storage area network (SAN) or a combination thereof. Such astorage device may access a device performing an embodiment of thepresent disclosure, via an external port. Further, a separate storagedevice on the communication network may access the device performing anembodiment of the present disclosure.

In the above-described detailed embodiments of the present disclosure, acomponent included in the present disclosure is expressed in thesingular or the plural according to a presented detailed embodiment.However, the singular form or plural form is selected for convenience ofdescription suitable for the presented situation, and variousembodiments of the present disclosure are not limited to a singleelement or multiple elements thereof. Further, either multiple elementsexpressed in the description may be configured into a single element ora single element in the description may be configured into multipleelements.

Although the present disclosure has been described with an exemplaryembodiment, various changes and modifications may be suggested to oneskilled in the art. It is intended that the present disclosure encompasssuch changes and modifications as fall within the scope of the appendedclaims.

Although the present disclosure has been described with an exemplaryembodiment, various changes and modifications may be suggested to oneskilled in the art. It is intended that the present disclosure encompasssuch changes and modifications as fall within the scope of the appendedclaims.

What is claimed is:
 1. An analog phase shifter comprising: a processor;a multi-port network comprising an input port for inputting a radiofrequency (RF) signal and an output port for outputting a phase-changedRF signal; a hybrid coupler configured to operably couple the input portand the output port to a plurality of load ports; and variable loadscoupled to the hybrid coupler via the plurality of load ports, whereinthe processor is configured to: identify a combination of controlvoltage values that is different from at least one combination ofcontrol voltage values causing an inner resonance of the variable loads;and apply a plurality of independent voltages according to theidentified combination of control voltage values to the variable loads.2. The analog phase shifter of claim 1, further comprising: a pluralityof direct current (DC) ports to which the plurality of independentvoltages are applied, wherein the variable loads comprise tunablereflective loads, wherein one of the tunable reflective loads comprisetwo varactors separated by a quarter (¼) wavelength transmission line,wherein the two varactors are connected to tuning open circuit lines toform varactor sections, respectively, wherein the plurality of DC portsare connected to the tuning open circuit lines via DC-pass filters, andwherein the quarter (¼) wavelength transmission line is grounded viaanother DC-pass filter.
 3. The analog phase shifter of claim 2, whereinone of the tuning open circuit lines has a length and a characteristicimpedance, wherein the length and the characteristic impedance aredetermined to achieve a phase shift of 180 degrees from a correspondingvaractor section for the one of the tuning open circuit lines.
 4. Theanalog phase shifter of claim 3, wherein the length and characteristicimpedance is based on an equation:${\left( {{- \frac{1}{2\;\pi\; f_{0}C_{\min}}} + X_{s}} \right)/\left( {2Z_{0}} \right)} = {{- 2}{Z_{0}/\left( {{- \frac{1}{2\;\pi\; f_{0}C_{\max}}} + X_{s}} \right)}}$wherein X_(s) is a full reactance of the one of the tuning open circuitlines having a parasitic inductance of a corresponding varactor, Z₀ isthe characteristic impedance, f₀ is a central frequency, C_(max) is amaximal value of a tunable capacitance of the corresponding varactor,and C_(min) is a minimal value of the tunable capacitance of thecorresponding varactor.
 5. The analog phase shifter of claim 4, whereinthe length of the one of the tuning open circuit lines is between onequarter (¼) and one half (½) of a transmission line wavelength at thecentral frequency f₀.
 6. The analog phase shifter of claim 1, furthercomprising: a plurality of direct current (DC) ports to which theplurality of independent voltages are applied, wherein the variableloads comprise tunable reflective loads, wherein one of the tunablereflective loads comprises two varactors separated by a quarter (¼)wavelength transmission line, wherein the two varactors comprise similarare connected to tuning shorted lines connected to form varactorsections, respectively, wherein the plurality of DC ports are connectedto the varactors via DC-pass filters, and wherein the quarter (¼)wavelength transmission line is connected in series with a blockingcapacitor.
 7. The analog phase shifter of claim 6, wherein one of thetuning shorted lines has a length and a characteristic impedance,wherein the length and the characteristic impedance are determined toachieve a phase shift of 180 degrees from a corresponding varactorsection for the one of the tuning shorted lines.
 8. The analog phaseshifter of claim 7, wherein the length and characteristic impedance isbased on an equation:${\left( {{- \frac{1}{2\;\pi\; f_{0}C_{\min}}} + X_{s}} \right)/\left( {2Z_{0}} \right)} = {{- 2}{Z_{0}/\left( {{- \frac{1}{2\;\pi\; f_{0}C_{\max}}} + X_{s}} \right)}}$wherein X_(s) is a full reactance of the one of the tuning shorted linehaving a parasitic inductance of a corresponding varactor, Z₀ is thecharacteristic impedance, f₀ is a central frequency, C_(max) is amaximal value of a tunable capacitance of the corresponding varactor,and C_(min) is a minimal value of the tunable capacitance of thecorresponding varactor.
 9. The analog phase shifter of claim 8, whereina reactance at a microwave frequency of the blocking capacitor is atleast an order of a magnitude less than Z₀.
 10. The analog phase shifterof claim 6, wherein one of the tunable reflective loads furthercomprise: an equalizing resistor connected in parallel with one varactoramong the two varactors; and a matching open circuit line connected inparallel with another varactor among the two varactors, wherein theequalizing resistor is further connected in series with another blockingcapacitor.
 11. The analog phase shifter of claim 1, further comprising:a memory configured to store information on a relationship between eachof a plurality of combinations of control voltage values and a phasechange amount of the RF signal, wherein the processor is, to identifythe combination of control voltage values, configured to: obtain thephase change amount of the RF signal; and identify the combination ofcontrol voltage values corresponding to the phase change amount based onthe information among the plurality of combinations of control voltagevalues.
 12. The analog phase shifter of claim 11, wherein the pluralityof the combinations of control voltage values are defined in a regionother than a region comprising the at least one combination of voltagevalues causing the inner resonance of the variable loads.
 13. Anoperating method of an analog phase shifter, the method comprising:identifying a combination of control voltage values that is differentfrom at least one combination of control voltage values causing an innerresonance of variable loads; and applying a plurality of independentvoltages according to the identified combination of control voltagevalues to the variable loads, wherein the applied plurality ofindependent voltages is used to output a phase-changed radio frequency(RF) signal from an RF signal.
 14. The method of claim 13, wherein theidentifying of the combination of control voltage values comprises:obtaining a phase change amount of the RF signal to output thephase-changed RF signal; and identifying the combination of the controlvoltage values corresponding to the phase change amount among aplurality of combinations of control voltage values, based oninformation on a relationship between each of a plurality ofcombinations of control voltage values and an amount of a phase change.15. The method of claim 14, wherein the plurality of combinations ofcontrol voltage values are defined in a region other than a regioncomprising the at least one combination of control voltage valuescausing the inner resonance of the variable loads.
 16. The method ofclaim 13, wherein each of the plurality of independent voltages isapplied to the variable loads through an independent control channel.17. The method of claim 16, wherein the plurality of voltages areapplied to a digital to analog converter (DAC) connected to controlports of the analog phase shifter.